Phase error estimation method for a demodulator in an HDTV receiver

ABSTRACT

A receiver for processing a VSB modulated signal containing terrestrial broadcast high definition television information and a pilot component includes a carrier recovery network ( 22 ; FIG.  3 ) that produces a demodulated baseband signal. The carrier recovery network additionally responds to a locally generated control signal (Ph. Offset;  360 ) representing an unwanted phase offset of the pilot signal due to multipath distortion, for example. The control signal is used to compensate for the pilot phase offset before the demodulated signal is equalized. The control signal is produced by correlating received sync values with both a reference sync value ( 362 ) and a Hilbert transform of the reference sync value ( 363 ). The output of the carrier recovery network signal is phase compensated twice.

FIELD OF THE INVENTION

This invention concerns a carrier recovery network for demodulating ahigh definition television signal, e.g., of the VSB-modulated typeadopted for use in the United States.

BACKGROUND OF THE INVENTION

The recovery of data from modulated signals conveying digitalinformation in symbol form usually requires three functions at areceiver: timing recovery for symbol synchronization, carrier recovery(frequency demodulation to baseband), and channel equalization. Timingrecovery is a process by which a receiver clock (timebase) issynchronized to a transmitter clock. This permits a received signal tobe sampled at optimum points in time to reduce slicing errors associatedwith decision-directed processing of received symbol values. Carrierrecovery is a process by which a received RF signal, after beingfrequency down converted to a lower intermediate frequency passband(e.g., near baseband), is frequency shifted to baseband to permitrecovery of the modulating baseband information. Adaptive channelequalization is a process by which the effects of changing conditionsand disturbances in the signal transmission channel are compensated for.This process typically employs filters that remove amplitude and phasedistortions resulting from frequency dependent time variantcharacteristics of the transmission channel.

SUMMARY OF THE INVENTION

In accordance with the principles of the present invention, a carrierrecovery network produces a demodulated signal in response to a pilotcarrier component of the received signal and a locally generated phasecorrection control signal representing an unwanted phase offset of thepilot signal transmitted with the main data signal. The control signalis a function of a correlation between a predetermined component of thereceived signal, and a reference value.

In an illustrated preferred embodiment, the control signal is producedby correlating a received sync component with both (a) a reference syncvalue, and (b) a Hilbert transform of the reference sync value.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a block diagram of a portion of a high definition television(HDTV) receiver including apparatus according to the principles of thepresent invention.

FIG. 2 depicts a data frame format for a VSB modulated signal employingthe ATSC high definition system in the United States.

FIG. 3 shows details of a carrier recovery demodulator network in FIG. 1in accordance with the present invention.

FIG. 4 is a diagram helpful in understanding the operation of thedemodulator in FIG. 1.

FIG. 5 shows additional details of a portion of the demodulator networkin FIG. 1.

DETAILED DESCRIPTION OF THE DRAWINGS

In FIG. 1, a terrestrial broadcast analog Input HDTV signal is processedby an input network 14 including RF tuning circuits and an intermediatefrequency (IF) processor 16 including a tuner for producing an IFpassband output signal, and appropriate automatic gain control (AGC)circuits. The received signal is a carrier suppressed 8-VSB modulatedsignal as proposed by the Grand Alliance and adopted as the ATSCterrestrial broadcast high definition television standard for use in theUnited States. Such a VSB signal is represented by a one-dimensionaldata symbol constellation wherein only one axis contains quantized datato be recovered by the receiver. To simplify the Figure, not shown aresignals for clocking the illustrated functional blocks.

As described in the Grand Alliance HDTV System Specification dated Apr.14, 1994, the VSB transmission system conveys data with a prescribeddata frame format as shown in FIG. 2. A small pilot carrier component(pilot tone) at the suppressed carrier frequency is added to thetransmitted signal to help a demodulator achieve carrier lock in a VSBreceiver. Referring to FIG. 2, each data frame comprises two fields witheach field including 313 segments of 832 multilevel symbols. The firstsegment of each field is referred to as a field sync segment, and theremaining 312 segments are referred to as data segments. The datasegments typically contain MPEG compatible data packets. Each datasegment comprises a four-symbol segment sync component followed by 828data symbols. Each field segment includes a four symbol segment synccharacter followed by a field sync component comprising a predetermined511 symbol pseudorandom number (PN) sequence and three predetermined 63symbol PN sequences, the middle one of which is inverted in successivefields. A VSB mode control signal (defining the VSB symbol constellationsize) follows the last 63 PN sequence, which is followed by 96 reservedsymbols and 12 symbols copied from the previous field. In the ATSCsystem, a small digital level (1.25) is added to every symbol (data andsyncs) of the digital baseband data plus sync signal. This has theeffect of adding a small in-phase pilot carrier component to the datasignal. Digital addition of the pilot at baseband provides a highlystable and accurate pilot. The frequency of the pilot is the same as thesuppressed carrier frequency.

Continuing with FIG. 1, the passband IF output signal from unit 16 isconverted to a digital symbol datastream by an analog to digitalconverter 19. The output digital datastream from ADC 19 is demodulatedto baseband by a digital demodulator/carrier recovery network 22. Thisis achieved by a phase locked loop in response to the pilot component inthe received VSB datastream. Unit 22 produces an output I-phasedemodulated symbol datastream as described in greater detail with regardto FIG. 3.

ADC 19 samples the input VSB symbol datastream in response to a samplingclock CLK. Associated with ADC 19 and demodulator 22 is a segment syncand symbol clock recovery network 24. Network 24 recovers the repetitivedata segment sync components of each data frame from the random data.The segment sync components are used to regenerate a properly phasedsampling clock.

Unit 28 detects the data field sync component by comparing everyreceived data segment with an ideal field reference signal stored inmemory in the receiver. In addition to field synchronization, the fieldsync signal provides a training signal for channel equalizer 34.Co-channel NTSC interference detection and rejection are performed byunit 30. Afterwards, the signal is adaptively equalized by channelequalizer 34 which may operate in a combination of blind, training, anddecision-directed modes. Equalizer 34 may be of the type described inthe Grand Alliance HDTV System Specification and in an article by W.Bretl et al., “VSB Modem Subsystem Design for Grand Alliance DigitalTelevision Receivers,” IEEE Transactions on Consumer Electronics, August1995. Equalizer 34 also may be of the type described in copending U.S.patent application Ser. No. 102,885 of Shiue et al. filed Jun. 23, 1998.

Equalizer 34 compensates for channel distortions, but phase noiserandomly rotates the symbol constellation. Phase tracking network 36removes any residual phase and gain noise present in the output signalfrom equalizer 34. The phase corrected signal is then trellis decoded byunit 40, de-interleaved by unit 42, Reed-Solomon error corrected by unit44, and descrambled (de-randomized) by unit-46. Afterwards, a decodeddatastream is subjected to audio, video and display processing by unit50.

Demodulation in unit 22 is performed by a digital automatic phasecontrol (APC) loop to achieve carrier recovery. The phase locked loopuses the pilot component as a reference for initial acquisition, anduses a conventional phase detector for phase acquisition. The pilotsignal is embedded in the received datastream, which contains dataexhibiting a random, noise-like pattern. The random data is essentiallydisregarded by the filtering action of the demodulator APC loop. Theinput signal to ADC 19 is a near baseband signal with the center of theVSB frequency spectrum at 5.38 MHz and the pilot component situated at2.69 MHz. In the demodulated datastream from unit 22 the pilot componenthas been frequency shifted down to DC.

FIG. 3 show details of digital demodulator 22. Demodulator 22 includes afirst phase control network 320, a second phase control network 350, anda phase correction signal generator 360. The operation of network 320will be described first.

The 8-VSB modulated digital symbol datastream from ADC 19, containingthe very low frequency pilot component, is applied to Hilbert filter 315that separates the incoming IF sampled datastream into mutuallyquadrature phased components “I” (in phase) and “Q” (quadrature phase).The I and Q components are rotated to baseband using complex multiplier324 in an automatic phase control (APC) loop. Once the loop issynchronized, the output of multiplier 324 is a complex baseband signalthat is further phase adjusted by network 350, as will be discussed, toproduce a final phase corrected demodulated output from unit 350. Theoutput I datastream from multiplier 324 is used to extract the pilotcomponent of the received datastream. The output Q datastream frommultiplier 324 is used to extract the phase of the received signal.

In the phase control loop, The Q signal is filtered by an automaticfrequency control (AFC) filter 336. High frequency data (as well asnoise and interference) are largely rejected by the AFC filter, leavingonly the pilot frequency. After filtering, the Q signal is amplitudelimited by unit 338 to reduce the dynamic range requirements of phasedetector 340. Phase detector 340 detects and corrects the phasedifference between the I and Q signals applied to its inputs, anddevelops an output phase error signal which is filtered by an APC filter344, e.g., a second order low pass filter. The phase error detected byunit 340 represents a frequency difference between the expected pilotsignal frequency near DC, and the received pilot component frequency.

If the received pilot component exhibits an expected frequency near DC,AFC unit 336 will produce no phase shift. The I and Q channel pilotcomponents input to phase detector 340 will exhibit no deviation from amutually quadrature phase relationship, whereby phase detector 340produces a zero or near zero value phase error output signal. However,if the received pilot component exhibits an incorrect frequency, AFCunit 336 will produce a phase shift. This will result in an additionalphase difference between the I and Q channel pilot components applied tothe inputs of phase detector 340. Detector 340 produces an output errorvalue in response to this phase difference.

The filtered phase error signal from filter 344 is provided tonumerically controlled oscillator (NCO) 348, which locally regeneratesthe pilot component for demodulating the received datastream. Associatedwith NCO 348 are sine and cosine look-up tables 349 for regenerating thepilot tone in response to the phase control signal from units 340 and344. The outputs of unit 349 are controlled until the I and Q signaloutputs of multiplier 324 cause the phase error signal produced bydetector 340 to be substantially zero, thereby indicating that ademodulated baseband I signal is present at the output of multiplier324.

As noted above, the pilot component in a received VSB modulated signalis tracked by a frequency and phase locked loop (FPLL), and therecovered pilot is used to heterodyne the received spectrum down tobaseband. When multipath, or “ghost,” components are present in thereceived spectrum, the carrier tracked by the phase locked loop is theresultant tone produced by the addition of the main path carriercomponent and the multipath component. This is illustrated in thediagram of FIG. 4. As shown in FIG. 4, multipath distortion produces anphase offset, or Phase Tracking Error, in the pilot, so that the pilotdoes not exhibit the correct demodulation phase with respect to thedata. Thus the reference pilot used for heterodyning has a phase offsetwith respect to the carrier in the main path, whereby the baseband mainpath signal receives a phase rotation through the heterodyning process.A subsequent channel equalizer, such as unit 34 in FIG. 1, may be ableto compensate for the effects of the pilot phase offset. However, thisoffset may cause the equalizer to use an excessively large amount of itsdynamic range to correct the pilot phase offset, or it may cause theequalizer to become unstable. The additional burden created by the pilotphase offset is removed by a method and apparatus according to a featureof the invention.

Networks 350 and 360 in FIG. 3 address the problem of the pilot phaseerror. Specifically, second phase control network 350 includes anadditional phase rotation network (multiplier) which can rotate therecovered signal independent of the pilot phase. This allows the pilotphase offset to be removed from the recovered data before the data isprocessed by equalizer 34. The equalizer therefore does not have tocompensate for the pilot phase offset, which permits the use of a lesscomplex equalizer design than would otherwise be needed. Phasecorrection signal generator 360 produces a Phase Offset control signalthat is used by phase control network 350 to compensate for the pilotphase offset.

In the illustrated preferred embodiment, the carrier recovery networkuses two rotators (multipliers) 324 and 356, both responsive to receivedI, Q signals. Rotator 324 is associated with a phase control loop innetwork 320 that responds to the pilot component. The other rotator,unit 356, is associated with control network 350 that additionallyresponds to a combined signal produced by combining a signal derivedfrom the phase control loop of network 320 with a Phase Offset controlsignal representing an estimate of undesired phase distortion, such asmultipath (“ghost”) distortion in the pilot signal. Network 360 producesthe Phase Offset control signal by correlating received segment syncvalues with both reference segment sync values and with a Hilberttransform of the reference segment sync value.

More specifically, multiplier 356 in network 350 receives the mutuallyquadrature phased I and Q signals from filter 315. Network 350 alsoreceives an input from the output of oscillator 348 in the phase lockedloop of network 320. This signal is combined in adder 352 with the PhaseOffset control signal produced by network 360 to compensate for thephase offset in the pilot carrier. The output signal of adder 352 is aphase compensated signal that is applied to look-up table 354 forproviding mutually quadrature phased output signals to complexmultiplier 356 (a second rotator). Look-up table 354 and multiplier 356operate in the same manner as look-up table 349 and complex multiplier(rotator) 324 in network 320. Multiplier 356 provides I and Q phasedoutput signals. The “I” phased output signal, compensated for the phaseoffset in the received pilot carrier, is applied to units 24 and 28, andeventually to equalizer 34, as shown in FIG. 1. Since any multipathinduced phase offset in the pilot carrier has been significantly reducedor eliminated by the coaction of networks 320, 350 and 360, theequalizer advantageously need not compensate for such offset. The secondoutput of complex multiplier 356, at which a “Q” phased signal wouldappear, is not used in this example.

The ATSC digital television modulation scheme employs a data field/frameformat as explained in connection with FIG. 2. Each data frame iscomposed of two data fields separated by a field sync component. Eachconstituent data field comprises a plurality of data segments eachprefaced by a segment sync component. These sync components occupyknown, fixed locations in the datastream, and will be referred to assync or sync components in the following discussion. After the receivedVSB modulated datastream has been demodulated to baseband and the synccomponents recovered (their locations identified), network 360 performsa correlation between the recovered segment sync components and bothknown segment sync amplitude values and the Hilbert transform of theknown segment sync amplitude values. The Hilbert transform produces aquadrature phased version of an applied input signal, as known.Correlated values are processed to obtain the Phase Offset controlsignal as follows. The field sync component and its transform can alsobe used by the correlation function.

Network 360 comprises first and second input correlators 362 and 363,both of which receive as inputs the received baseband segment syncsamples. Correlator 362 additionally receives constant segment syncvalue “S” from local memory, and correlator 363 additionally receives aHilbert transformed constant segment sync value “H(S)” from localmemory. The correlation produced by unit 362 produces an output value Icdefined by the following expression

Ic=Gc|S|² cos Φ

where |S|² is the result of correlating the known sync component withitself, and Gc is an arbitrary gain factor. The correlation produced byunit 363 produces an output value Is defined by the expression

Is=Gs|H(S)|² sin Φ

where |H(S)|² is the result of correlating the Hilbert transform of theknown sync component with itself, and Gs=Gc. The pilot phase trackingerror is designated by the symbol Φ.

The Is and Ic outputs of correlators 362 and 363 are processed by anetwork 365 which produces the mathematical value Is/Ic, or

Is/Ic=|H(S)|²/|S|²×(sin Φ)/(cos Φ

The values of the expressions |H(S)|² and |S|² are known since they arefunctions of known sync component values. The term |H(S)|²/|S|² in theabove expression is cancelled by the multiplication of this term withits inverse (a stored constant) in multiplier 366, producing thefollowing expression at the output of multiplier 366

Is/Ic=sin Φ/cos Φ=tan Φ, so that

Φ=tan⁻¹(Is/Ic).

The term Is/Ic is a numerical value which is used in tan⁻¹ look-up table367 to determine the value of offset phase shift Φ. The output valuefrom look-up table 367 is applied to a “D” input of latch 368, e.g., aD-type flip-flop. An enable input EN of latch 368 receives a locallygenerated Sync Position Detected signal when segment sync has beenrecovered by timing recovery unit 24 (FIG. 1). The Sync PositionDetected signal is provided in this example by unit 24, although a localmicroprocessor that monitors the operations of segment sync timingrecovery network 24 could also provide this signal. The Sync PositionDetected signal enables latch 367 to output the phase offset signalreceived at its D input to network 350 as the Phase Offset controlsignal for use as discussed above.

The demodulated I channel datastream from network 350 is applied tosegment sync and symbol clock recovery unit 24 and to field syncdetector 28 as shown in FIG. 1. When the repetitive data segment syncpulses are recovered from the random data pattern of the receiveddatastream, the segment syncs are used to achieve proper symbol timingby regenerating a properly phased symbol sampling clock.

Following is a more detailed description of the operation of pilot phaseoffset estimator network 360. The input signal applied to correlators362 and 363 is of the form I(n)=x(n) cos Φ−x′(n) sin Φ, where x′ is theHilbert transform of x(n), and the pilot carrier phase offset error tobe corrected is Φ. The multi-symbol segment sync pattern for the ATSCsystem is designated as S, and its Hilbert transform is designated as H′(or H(s)) as discussed above). Correlating S and S′ yields Ic=|S|²cos Φand Is=−|S′|²sin Φ. Since S and S′ are constants, it can be seen thatIs/Ic is proportional to −Ctan Φ where C is a constant. For values of Φe.g., between −90 and +90 degrees, tan Φ approaches Φ so that Φ isapproximately equal to −(Is/Ic)×(1/C). For small values of Φ Ic isgreater than zero, so neglecting the factor Ic×C results in Φapproximately equal to some positive scaling of −Is.

Since the four symbol segment sync pattern normally has symbol values+160−160−160+160, normalizing S′ to +1 and −1 values producesS′=+1−1−1+1 corresponding to the normal segment sync pattern. Thisnormalization simplifies the Is, Ic correlation process to a process ofaddition whereby the phase offset error Φe is approximately equal toΦe=−S0−S1+S2+S3 where S0, S1, S2 and S3 represent the four symbolsconstituting a segment sync pattern. In order to reduce the impact ofnoise and multipath on the calculation of Φe, the individual syncsymbols S0, S1, S2, S3 are each averaged over a predetermined interval Tencompassing 64 consecutive segment sync patterns, for example.Afterwards, a correlation value e′ is produced according to theexpression$e^{\prime} = \frac{{- {S0}^{\prime}} - {S1}^{\prime} + {S2}^{\prime} + {S3}^{\prime}}{T}$

This value is summed over time (e.g., 64 segment sync intervals) andscaled by a predetermined scale factor G to produce a final estimatevalue “e.” Scale factor G is determined empirically and sets thetracking bandwidth.

The process described above is illustrated by the arrangement shown inct; FIG. 5. In FIG. 5 an input adder network is constituted by units512, 513, 514, 515, 525 and 528 arranged as shown. Associated with theadder network are delay elements 518, 519, 520 and 521. Each delayelement represents a one symbol delay. The output of adder 525 issubtractively combined with the output of adder 528 in unit 530. Theoutput of combiner 530 is processed by unit 532 to produce the value e′in accordance with the above expression. Unit 532 divides by the numberof segment sync components (64) that were summed during previousprocessing, thereby producing an expected sync amplitude. Afterwardsvalue e′ is processed by adder 534 and associated symbol delay 535, andscaled by unit 538 to produce the final error estimate e. The inputadder network is reset at the end of each interval T.

What is claimed is:
 1. In a receiver for processing a received VestigialSideband (VSB) modulated signal containing a digital pilot carriercomponent and digital video data represented by a VSB symbolconstellation, said data having a data frame format constituted by asuccession of data frames comprising a field a sync component prefacinga plurality of data segments each having an associated segment sync, amethod comprising the steps of: generating a control signal representinga phase offset error of said pilot component; and demodulating saidreceived signal in response to the phase of said pilot component and tosaid control signal concurrently; wherein said generating step includesthe step of: generating the control signal as a function of acorrelation between a value of a predetermined component of the receivedsignal, and a reference value.
 2. A method according to claim 1, whereinsaid predetermined component is a segment sync component of the receivedsignal.
 3. In a receiver for processing a received Vestigial Sideband(VSB) modulated datastream containing a pilot carrier component andvideo data represented by a VSB symbol constellation, said data having adata frame format constituted by a succession of data frames comprisinga field of sync component prefacing a plurality of data segments eachhaving an associated segment sync, a method comprising the steps of:generating a control signal representing a phase offset error of saidpilot component; and demodulating said received signal in response tosaid pilot component and to said control signal: wherein said generatingstep includes the step of: generating the control signal as a functionof a correlation between a value of a predetermined component of thereceived signal, and a reference value; wherein said correlationcomprises the steps of correlating a received sync value with areference sync value; and correlating said received sync value with atransformed reference sync value.
 4. A method according to claim 3,wherein said transformed reference sync value is a Hilbert transformedvalue.
 5. A method according to claim 3, wherein said sync is a segmentsync component of the received signal.
 6. A method according to claim 3,wherein said sync is a segment sync component of the received signal;and said generating step comprises the steps of correlating a receivedsegment sync value with a reference segment sync value; and correlatingsaid received segment sync value with a transformed reference segmentsync value.
 7. A method according to claim 6, wherein the transformedvalue is a Hilbert transformed value.